Delay compensation circuit

ABSTRACT

A device ( 200 ) includes a circuit ( 202 ) and a driver stage ( 204 ) therefor. The circuit includes two sub-circuits ( 231  and  232 ). The driver stage includes switcher logic ( 206 ) that produces signals that control switching on and off of the sub-circuits. The switcher logic also produces other signals in advance of the signals that control the switching of the sub-circuits. The driver stage includes delay compensations circuits ( 221  and  222 ), coupled to the switcher logic and to the circuit, that produce timing signals for the switcher logic. The timing signals are closely aligned with moments that a changing voltage at a node between the sub-circuits passes through threshold voltages. The timing signals compensate for all delays of signals through the device such that a period that both sub-circuits are off is minimized, while ensuring that both sub-circuits are not on at a same time.

BACKGROUND

1. Field

This invention relates generally to delay compensation circuits, andmore specifically a delay compensation circuit that can be used with adriver stage of a switched-mode power supply.

2. Related Art

A switched-mode power supply (hereinafter “SMPS”) is a type of directcurrent voltage to direct current voltage converter that includes twopower transistors in a push-pull configuration and an inductor coupledbetween the two power transistors and an output terminal of the SMPS.

A driver stage of a SMPS alternately switches the two power transistorson and off. To ensure that only one of the two power transistors are onat any moment, the driver stage creates a dead time that occurs duringswitching transitions. Dead time is defined as an interval during whichboth power transistors are off. The dead time occurs at each cycleduring a period between when one power transistor turns off and theother power transistor turns on. The presence of the dead time avoids asituation where both power transistors are on simultaneously, whichwould cause a short circuit current and may also lead to failure of thepower transistors. The dead time causes an output voltage of the SMPS tobe clamped around −0.6V (one diode voltage drop) due to the effect ofthe inductor. The dead time inserts losses into the SMPS, and suchlosses become more significant when the output voltage of the SMPS isrelatively low. For example, if a SMPS has an output voltage above 4V,the diode voltage drop during the dead time may not be a problem, but ifa SMPS has an output voltage of 1.8V or less, the diode voltage dropduring the dead time may compromise efficiency of the SMPS. A SMPShaving an output voltage of 1.8V or less is used as a power supply forsome microprocessors.

A high switching frequency of a SMPS allows components, such asinductors and capacitors, to be small in size and allows the SMPS tohave a fast reaction time relative to changes in load current.Disadvantageously, the losses due to the dead time become moresignificant as the switching frequency is increased.

FIG. 1 illustrates a known device 100 comprising an output stage of aSMPS 102 coupled to a known driver stage 104. The known driver stage 104comprises known switcher logic 106, an upper buffer 108, a lower buffer110, a first comparator 121 and a second comparator 122. The firstcomparator 121 outputs, to the known switcher logic 106, an F_LATCHAsignal. The second comparator 122 outputs, to the known switcher 106, anR_LATCHA signal. The SMPS 102 comprises an upper power transistor 131, alower power transistor 132, an inductor 134 coupled between the powertransistors, and an output terminal 140 of the SMPS, as illustrated inFIG. 1. The SMPS 102 also comprises diodes 135 and 136 that areinherently within power transistors 131 and 132, respectively. A loadcapacitor 137 and a load resistor 138 are coupled to the output terminal140 of the SMPS 102.

A variable voltage SWA appears at a node 133 between the two powertransistors 131 and 132. The inductor 134 is coupled between node 133and the output terminal 140 of the SMPS 102. The inductor 134 affectsthe voltage at node 133. The first comparator 121 compares the SWAvoltage with a fixed DC reference voltage V_(REF), which is typicallychosen to be slightly above 0V. The second comparator 122 compares theSWA voltage with a fixed DC reference voltage V_(REFR), which istypically chosen to be slightly below 0V.

The known switcher logic 106 of the driver stage 104 generates W_VGUAand W_VGLA signals that are fed into the upper buffer 108 and the lowerbuffer 110, respectively, of the SMPS 102. The known switcher logic 106generates the W_VGUA and W_VGLA signals based, in part, upon timing ofthe F_LATCHA signal from the first comparator 121 and the R_LATCHAsignal from the second comparator 122.

The upper buffer 108 and the lower buffer 110 output VGUA and VGLAsignals, respectively, that alternately turn on and off the two powertransistors 131 and 132 such that only one power transistor is on at anymoment. The VGUA signal alternately turns on and off the upper powertransistor 131. The VGLA signal alternately turns on and off the lowerpower transistor 132.

Disadvantageously, the known driver stage 104 may be unable to reducethe dead time sufficiently enough for optimal efficiency.

Known driver stages, such as known driver stage 104, may increase thedead time when inherent delays associated with comparators and logicpaths have a longer duration than the time of one sweep of the SWAvoltage.

Furthermore, inherent delays associated with comparators and logic pathsmay vary with process, noise, temperature, V_(DD) variation, clock andtiming variation, part-to-part variation, and other random effects. Thedead time of known driver stages, including known driver stage 104, mayincrease when such inherent delays are unknown and/or vary.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention is illustrated by way of example and is notlimited by the accompanying figures, in which like references indicatesimilar elements. Elements in the figures are illustrated for simplicityand clarity and have not necessarily been drawn to scale.

FIG. 1 is a simplified functional block diagram of a known deviceincluding an output stage of a switched-mode power supply and a knowndriver stage therefor.

FIG. 2 is a simplified functional block diagram of a device inaccordance with one embodiment of the invention, including an outputstage of a switched-mode power supply and one embodiment of a driverstage therefor which includes switcher logic and a delay compensationcircuit.

FIG. 3 is a logic diagram of one embodiment of the switcher logic of thedriver stage illustrated in FIG. 2.

FIG. 4 is a schematic and block diagram of one embodiment of the delaycompensation circuit of the driver stage illustrated in FIG. 2, whichincludes a sampler circuit and a series resistance.

FIG. 5 is a graph of exemplary signals that may be present in the deviceillustrated in FIG. 2, in accordance with one embodiment of theinvention.

FIG. 6 is a pair of graphs illustrating results of operation of asimulation of the known device illustrated in FIG. 1.

FIG. 7 is a pair of graphs illustrating results of operation of asimulation of the device illustrated in FIG. 2, in accordance with oneembodiment of the invention.

FIG. 8 is a schematic diagram of one embodiment of a circuit thatemulates the sampler circuit and the series resistance of the delaycompensation circuit illustrated in FIG. 4.

DETAILED DESCRIPTION

FIG. 2 illustrates a device 200 in accordance with one embodiment of theinvention, which comprises an output stage of a switched-mode powersupply (hereinafter “SMPS”) 202 coupled to a driver stage 204 inaccordance with the invention. The driver stage 204 comprises switcherlogic 206 in accordance with the invention, an upper buffer 208, a lowerbuffer 210, a first delay compensation circuit 221 in accordance withthe invention and a second delay compensation circuit 222 in accordancewith the invention. In one embodiment, the driver stage 204, the firstdelay compensation circuit 221 and the second delay compensation circuit222 are disposed on a circuit-supporting substrate of an integratedcircuit. The upper buffer 208 and the lower buffer 210 may also bedisposed on the circuit-supporting substrate of the integrated circuit.

The driver stage 204 implements a threshold locked loop (TLL) circuit.Advantageously, the TLL circuit in accordance with one embodiment of theinvention can work with waveforms not suitable for the known driverstage 104, such as a waveform that has a very fast waveform sweep.Unlike known delay compensation circuits, the TLL circuit in accordancewith one embodiment of the invention can work with another circuit, suchas the output stage of the SMPS 202, that has a large delay for which tobe compensated (large compared to a sweep of a waveform in the othercircuit). Unlike known TLL circuits, the TLL circuit in accordance withone embodiment of the invention can work with another circuit, such asthe output stage of the SMPS 202, that has a waveform with non-monotonicbehavior close to a threshold point, V_(REF). The phrase “close to athreshold point” means within a time interval less than or equal to atotal delay within the device 200. When the voltage of the waveform isclose to V_(REF), the shape of the waveform should be the same for eachsuccessive cycle of the waveform. For example, if a waveform has arising rate of 1V/1 ns when the voltage of the waveform is close toV_(REF), then the waveform should always have the same rising rate of1V/1 ns for each successive cycle of the waveform when the voltage ofthe waveform is close to V_(REF). This condition ensures that a stablesteady state is achievable.

The first delay compensation circuit 221 outputs, to the switcher logic206, an F_LATCH signal. The second delay compensation circuit 222outputs, to the switcher logic 206, an R_LATCH signal. The SMPS 202comprises an upper power transistor 231, a lower power transistor 232,and an inductor 234 coupled between the power transistors, and an outputterminal 240 of the SMPS, as illustrated in FIG. 2. In one embodiment,the inductor 234 has an inductance of 4.7 μH. The SMPS 202 alsocomprises diodes 235 and 236 that are inherently within powertransistors 231 and 232, respectively. A load capacitor 237 and a loadresistor 238 are coupled to the output terminal 240 of the SMPS 202, asillustrated in FIG. 2. In one embodiment of the device 200, V_(DD)=3.3V,and V_(OUT)=1.2V. A variable voltage SW appears at a node 233 betweenthe two power transistors 231 and 232. In the one embodiment, the SWvoltage has a maximum voltage of 3.3V and a minimum voltage slightlybelow 0V. The inductor 234 is coupled between node 233 and the outputterminal 240 of the SMPS 202. The inductor 234 affects the SW voltage atnode 233. In one embodiment, the upper power transistor 231 is ann-channel metal-oxide-semiconductor field-effect transistor (hereinafter“MOSFET”) and the lower power transistor 232 is an n-channel MOSFET, andnode 233 is coupled to a source terminal of upper power transistor 231and to a drain terminal of lower power transistor 232.

Current through the inductor 234 changes with time directlyproportionally to the voltage across it and inversely proportional toits inductance, i.e., di_(L)/dt=V(t)/L. When the upper power transistor231 is on, the voltage across the inductor 234 is positive and thecurrent through the inductor ramps up. When the lower power transistor232 turns on, the voltage across the inductor 234 is negative and thecurrent through the inductor ramps down. In despite of the ramping upand the ramping down, the current through the inductor 234 is positivemost of the time and is sometimes zero, and, for good SMPS designs, thecurrent through the inductor never ramps to negative values. When theupper power transistor 231 turns off and the lower power transistor 232has not yet turned on, the current through the inductor 234 reaches itsmaximum value, and the inductor 234 makes the SW voltage go down inorder to decrease the current through the inductor. The decreasing SWvoltage cannot go below −0.6V because diode 236, which is inherentlywithin the lower power transistor 232, turns on when SW=−0.6V, which isone diode voltage drop. With the known driver stage 104, the SWA voltagedisadvantageously goes down to −0.6V during each dead time (see FIG. 6).The dead time is an interval during which both power transistors 231 and232 are off. On the other hand, with the driver stage 204 in accordancewith the invention, the SW voltage advantageously does not go down to−0.6V during the dead time, for the reason explained more fullyhereinbelow (see FIG. 7). For inductance values and fast SW voltagesweeps that are typically used in the SMPS 202, the current through theinductor 234 may be considered constant during any given sweep of the SWvoltage.

The switcher logic 206 of the driver stage 204 alternately generatesdigital W_VGU and W_VGL signals, which are fed into the upper buffer 208and the lower buffer 210, respectively, of the SMPS 202. The switcherlogic 206 is coupled to a clock circuit (not shown) that outputs a CLOCKsignal. The switcher logic 206 cyclically generates the W_VGU and W_VGLsignals at a frequency. The frequency of the W_VGU and W_VGL signalsdetermines a frequency that the SW voltage sweeps through 0V. In oneembodiment, the frequency that the SW voltage sweeps through 0V is 8MHz. The upper buffer 208 and the lower buffer 210 delay the W_VGU andW_VGL signals, respectively. The upper buffer 208 and the lower buffer210 also ensure that the analog VGU and VGL signals, respectively, haveproper power and voltage levels to turn on the power transistors 231 and232, respectively, at a required SMPS frequency. In one embodiment, suchproper level is 6V. The switcher logic 206 generates the W_VGU and W_VGLsignals based, in part, upon timing of the F_LATCH signal from the firstdelay compensation circuit 221 and the R_LATCH signal from the seconddelay compensation circuit 222. Unlike known switcher logics, such asswitcher logic 106, the switcher logic 206 also generates a digitalP_VGU signal for the first delay compensation circuit 221 and a digitalP_VGL signal for the second delay compensation circuit 222. The switcherlogic 206 advantageously generates the P_VGU and the P_VGL signalsbefore it generates the W_VGU and the W_VGL signals, respectively. TheP_VGU and P_VGL signals are digital signals that are generated such thatthey have a frequency equal to a frequency that the SW voltage sweepsthrough 0V. The P_VGU and P_VGL signals are generated such that theirfalling (in one embodiment) edge occurs a pre-selected amount of timeprior to the falling (in one embodiment) edge of the W_VGU and W_VGLsignals, respectively. The pre-selected amount of time is chosen suchthat it is greater than a total delay that is expected to occur withinthe device 200.

The upper buffer 208 is coupled to a control electrode of upper powertransistor 231, and the lower buffer 210 is coupled to a controlelectrode of lower power transistor 232. The upper buffer 208 and thelower buffer 210 output the analog VGU and VGL signals, respectively,that alternately turn on the two power transistors 231 and 232 such thatonly one power transistor is on at any moment. The VGU signalalternately turns on the upper power transistor 231. The VGL signalalternately turns on the lower power transistor 232. In one embodiment,the VGU and VGL signals are identical to the W_VGU and W_VGL signals,respectively, except that they are delayed in time. In anotherembodiment, the VGU and VGL signals are delayed in time and may havehigher power and/or voltage than the W_VGU and W_VGL signals in order toproperly turn on the power transistors 131 and 132.

It is desirable to align, as close as feasible, the turning on of thepower transistors 231 and 232 with a moment that the SW voltage sweepsthrough 0V. Such alignment ensures that the dead time has a minimalduration. See FIG. 7, which shows several cycles of the SW voltage ofthe device 200. A negative SW voltage exists during the dead time 701.Advantageously, the driver stage 204 aligns, very closely, the turningon of the power transistors 231 and 232 with the moment that the SWvoltage sweeps through 0V, thereby minimizing duration of the dead time701.

The portions of the driver stage 204 shown in FIG. 2 are involved in aproper timing of the turning on, as opposed to the turning off, of thepower transistors 231 and 232. The switcher logic 206 also causes thepower transistors 231 and 231 to turn off as the CLOCK signal changesits state. However, the turning off of the power transistors 231 and 232is performed in a straightforward manner because the turning off of thepower transistors is a safe action in that there is little danger ofboth power transistors becoming on at a same time due to improper timingof the turning off action because the VGL signal that turns on the lowerpower transistor 232 is initiated by a same action within the switcherlogic 206 that turns off of the upper power transistor 231, and viceversa.

FIG. 3 is a schematic diagram of one embodiment of the switcher logic206. The switcher logic 206 comprises an inverter 371 and an OR gate372, and an output of each inverter is coupled to a NOR gate 373. Anoutput of NOR gate 373 is fed into a delay element 374, an OR gate 375and into a first input terminal of an OR gate 376. An output of delayelement 374 is fed into a second input terminal of OR gate 376. Anoutput of OR gate 375 is fed into a NOR gate 377. An output of NOR gate377 is fed into a delay element 378, the OR gate 372 and into a firstterminal of an OR gate 379. An output of delay element 378 is fed into asecond input terminal of OR gate 379. One embodiment of the switcherlogic 206 receives the R_LATCH signal, the F_LATCH signal and the CLOCKsignal, as illustrated in FIG. 3. The OR gates 376 and 379 inconjunction with the delay elements 374 and 378 allow the switcher logic206 to generate later falling edges of the W_VGU and W_VGL signalsrelative to falling edges of the P_VGU and P_VGL signals, respectively.Therefore, transitions of the P_VGU and P_VGL signals occur earlier thantransitions of the W_VGU and W_VGL signals, respectively. In oneembodiment, delay elements 374 and 378 delay the signal through them byabout 10 ns to 40 ns. The switcher logic 206 may include additionalgates (not shown) for receiving an ENABLE signal. The one embodiment ofthe switcher logic 206 outputs the P_VGU signal, the P_VGL signal, theW_VGU signal and the W_VGL signal, as illustrated in FIG. 3.

A purpose of the first delay compensation circuit 221 is to controltiming of generation of the VGL signal which turns on the lower powertransistor 232 (after the upper power transistor 231 has been turnedoff). A goal of the first delay compensation circuit 221 is to turn onthe lower power transistor 232 as soon as possible after the upper powertransistor 231 is turned off. The switcher logic 206 generates the W_VGLsignal. As a result of such generation, the buffer 210 outputs the VGLsignal. In one embodiment, a high VGL signal turns on the lower powertransistor 232. The first delay compensation circuit 221 ensures that arising edge of the VGL signal occurs at a moment that a falling SWvoltage passes through a same voltage level as V_(REF) which is a fixedDC reference voltage. In general, a value of V_(REF) is chosen such thatit is between the minimum and maximum values of the SW voltage. In oneembodiment, V_(REF) is chosen such that it is as close to 0V as feasibleto ensure that the upper power transistor 231 is off when the falling SWvoltage reaches V_(REF). In one embodiment, V_(REF) is chosen such thatit is above 0V for design convenience. In one embodiment, V_(REF)=0.1V.The first delay compensation circuit 221 compensates for delays withinthe device 200 by outputting the F_LATCH signal at an appropriate momentso as to cause the lower power transistor 232 to turn on at a propermoment. The first delay compensation circuit 221 compensates for suchdelays in spite of the fact that a duration of the delays is unknown.

A purpose of the second delay compensation circuit 222 is to controltiming of generation of the VGU signal which turns on the upper powertransistor 231 (after the lower power transistor 232 has been turnedoff). A goal of the second delay compensation circuit 222 is to turn onthe upper power transistor 231 as soon as possible after the lower powertransistor 232 was turned off. The switcher logic 206 generates theW_VGU signal. As a result of such generation, the buffer 208 outputs theVGU signal. In one embodiment, a high VGU signal turns on the lowerpower transistor 232. The second delay compensation circuit 222 ensuresthat a rising edge of the VGU signal occurs at a moment that a fallingSW voltage (occurring immediately before a rising SW voltage) passesthrough a same voltage level as a V_(REFR) fixed DC reference voltage.See, for example, the falling SW voltage at 39.12 μs in FIG. 7. Ingeneral, a value of V_(REFR) is chosen such that it is between a minimumvalue of the SW voltage and 0V. In one embodiment, the value of V_(REFR)is chosen such that it is as close to 0V as feasible, but still below0V. In one embodiment, V_(REFR)=−0.1V. The second delay compensationcircuit 222 compensates for delays within the device 200 by outputtingthe R_LATCH signal at an appropriate moment so as to cause the upperpower transistor 231 to turn on at a proper moment. The second delaycompensation circuit 222 compensates for such delays in spite of thefact that the duration of the delays is unknown.

When the upper power transistor 231 is on, the SW voltage is equal toV_(DD) minus a drain-to-source voltage drop of the upper powertransistor. Assuming, in one embodiment, V_(DD)=3.3V. When the lowerpower transistor 232 is on, the SW voltage is equal to 3.27V becausethere is a 30 mV voltage drop in the upper power transistor 231 due itsinherent on resistance. The first delay compensation circuit 221 alignsoccurrence of the rising edge of the VGL signal with occurrence of acrossing of the constant reference voltage V_(REF) by the variable SWvoltage. The falling edge of the VGU signal occurs earlier in eachcycle, and it is the falling edge of the VGU signal that turns off theupper power transistor 231 and causes the SW voltage to fall. Thefalling edge of the VGU signal depends only on the CLOCK signal. As aresult of this alignment, the first delay compensation circuit 221compensates for a total delay within the device 200. In this context,“total delay” means a period between the moment than the SW voltageequals V_(REF) and the moment that power transistor 232 turns on. Thetotal delay includes a delay through the switcher logic 206 (typically,a few hundred of picoseconds), a delay through one of the buffers 208(typically, tens of nanoseconds), a delay through one of the powertransistors 231 (typically, 4 ns to 30 ns), and a delay through thefirst delay compensation circuit 221 itself (typically, 3 ns to 5 ns).In one embodiment of the device 200, the total delay is typically 20 ns.The first delay compensation circuit 221 accomplishes the compensationin spite of the fact that the SW voltage does not need to be a smoothvoltage function or have a monotonic voltage variation. On the otherhand, with the known driver stage 104, prior to occurrence of eachSWA=V_(REF) event, the SWA voltage must have a monotonic voltagevariation for a period equal to at least the total delay in the knowndevice 100.

When the upper power transistor 231 is turned off, the SW voltage fallsfrom V_(DD) minus a drain-to-source voltage drop toward −0.6V. A veryshort time after the upper power transistor 231 turns off, the lowerpower transistor 232 turns on, and the SW voltage advantageously doesnot reach −0.6V. When the lower power transistor 232 is on, the SWvoltage becomes −0.03V because the lower power transistor 232, when itis on, has a drop voltage caused by its on resistance. When the lowerpower transistor 232 is turned off, the SW voltage would begin to fallfrom −0.03V toward −0.6V.

Both the first delay compensation circuit 221 and the second delaycompensation circuit 222 receive the SW voltage as one of their inputs.The first compensation circuit 221 receives the P_VGU signal, the VGLsignal, and the V_(REF) voltage as additional inputs. The secondcompensation circuit 222 receives the P_VGL signal instead of the P_VGUsignal, the VGU signal instead of the VGL signal, and the V_(REFR)voltage instead of the V_(REF) voltage as additional inputs. The firstcompensation circuit 221 outputs the F_LATCH signal. The secondcompensation circuit 222 outputs the R_LATCH signal instead of theF_LATCH signal. In other respects, the first delay compensation circuit221 operates in essentially a same fashion as the second delaycompensation circuit 222. Because the first delay compensation circuit221 operates in essentially the same fashion as the second delaycompensation circuit 222, only the operation of the first delaycompensation circuit (hereinafter “compensation circuit”) 221 will bedescribed in detail.

The compensation circuit 221 accomplishes edge alignment control. Thephrase “edge alignment control” means that the compensation circuit 221aligns a rising edge of the VGL signal with a moment that SW=V_(REF)during a falling SW voltage, i.e., during a downward sweep of the SWvoltage.

FIG. 4 is a schematic and block diagram of one embodiment of thecompensation circuit 221. The compensation circuit 221 includes asampler circuit 401 that samples and holds the SW voltage at eachoccurrence of a rising edge of the VGL signal. In the illustratedembodiment, the sampler circuit 401 is sensitive to the rising edge ofthe VGL signal, such that, when the rising edge of the VGL signaloccurs, a value of the SW voltage at that moment is copied to a node410. In another embodiment (not shown) of the compensation circuit 221,a falling edge of the SW voltage is aligned with a falling edge of theVGL signal at the moment that SW=V_(REF) and, in such embodiment, adifferent sampler circuit is implemented. The compensation circuit 221,in the illustrated embodiment, aligns the SW voltage with a rising edgeof the VGL signal at the moment that the variable voltage SW equals theconstant reference voltage V_(REF).

The compensation circuit 221 includes an integrator 405. In theillustrated embodiment, the integrator 405 comprises a resistor 406, acapacitor 407 and an operational amplifier 408. An input node 411 of theamplifier 408 receives the V_(REF) signal, and an inverted input node409 of the amplifier is connected to capacitor 407 and resistor 406 toperform the integration. The integrator 405 integrates a voltagedifference between the SW voltage and V_(REF). A time constant of theintegrator 405 is chosen to be much larger than a period of the SWvoltage waveform. In one embodiment, the time constant is at least forty(40) times greater than the period of the SW voltage waveform. An outputof the integrator 405 is a VIN_AUX signal. Initially, the SW voltage atnode 410 is lower than the voltage V_(REF) at the input node 411 of theamplifier, and the output voltage VIN_AUX of the integrator increaseswith each sampling event because the integrator has an inversioncharacteristic. A rate of increase of VIN_AUX depends on values of theresistor 406 and capacitor 407 of the integrator 405. In accordance withthe invention, the rate of increase is chosen to be much slower than afrequency of the SW voltage waveform. In one embodiment, the rate ofincrease is chosen to be 15 mV/μs.

The compensation circuit 221 includes a waveform generator 420. TheP_VGU signal is fed into an input terminal of the waveform generator420. A falling (in one embodiment) edge of the P_VGU signal triggersgeneration of the VC_AUX waveform. The waveform generator 420 generatesa VC_AUX waveform which is a smooth and monotonically decreasingvoltage. In one embodiment, the VC_AUX waveform decreases from 1.2V to0.2V in 84 ns. The waveform generator 420 generates the VC_AUX waveformat start-up of the driver stage 204. The waveform generator 420 alsogenerates the VC_AUX waveform at the beginning of every cycle of the SWvoltage.

The compensation circuit 221 includes a comparator 430 that has oneinput terminal for receiving the VC_AUX signal, an inverted inputterminal for receiving the VIN_AUX signal, and an output terminal foroutputting a COMP_AUX signal.

The compensation circuit 221 also includes a comparator 440 that has oneinput terminal for receiving the SW voltage, an inverted input terminalfor receiving the V_(REF) signal, and an output terminal for outputtinga COMP_(—)0 signal. Comparator 440 is used during initialization andduring a forced fall back.

The compensation circuit 221 also includes an inverter 450 that receivesthe P_VGU signal and that outputs a P_VGU_B signal.

The compensation circuit 221 further includes an RS (reset-set) latch460 having input terminals for receiving the COMP_(—)0 signal, theCOMP_AUX signal and the P_VGU_B signal, and an output terminal foroutputting an F_LATCH signal. The RS latch 460 includes a three-inputNAND gate 465 and a two-input NAND gate 464 configured as illustrated inFIG. 4. The F_LATCH signal outputted by the RS latch 460 becomes thesignal outputted by the compensation circuit 221.

A threshold voltage of comparator 430 is generated by the VIN_AUXsignal, which is an error function. The VIN_AUX signal is generated byintegrator 405 from a difference between the reference voltage V_(REF)and the variable voltage SW that is sampled at a moment that the VGLsignal, which is a feedback signal, changes state. The compensationcircuit 221 produces a steady state F_LATCH signal, which is also afeedback signal. These feedback signals, and other signals in the device200, encounter delays. To reduce dead time, the compensation circuit 221compensates for such delays.

The operation of the compensation circuit 221 can be better understoodby first analyzing it when it is in an initial state, next analyzing itwhen it is in a middle state, and then analyzing it when it is in asynchronized state. The operation of the compensation circuit 221 canalso be appreciated by referring to FIG. 5. In the following explanationof the operation of compensation circuit 221, it is assumed that the SWvoltage is sweeping downwardly.

Initial state. By “initial state” it is meant the state of thecompensation circuit 221 in a first cycle of the SW voltage waveform. Inthe initial state, a VC_AUX=VIN_AUX event 509 occurs after theSW=V_(REF) event.

To understand the operation of the compensation circuit 221, assume thata voltage of output VIN_AUX of the integrator 405 is initially very low.Assume also that a falling edge of the P_VGU signal is generated by theswitcher logic 206 a certain amount of time prior to occurrence of thefalling edge of the SW voltage. As explained more fully below, thecompensation circuit 221 uses the P_VGU signal to help compensate forthe total delay that occurs within the device 200.

The COMP_AUX output of the comparator 430 goes low when VC_AUX equalsVIN_AUX. The voltage VIN_AUX is very low during the initial state. TheVC_AUX signal starts high, and, although it immediately starts to drop,it remains higher than VIN_AUX during the initial state. As a result,the COMP_AUX output of the comparator 430 starts high at start-up of thecompensation circuit 221 and goes low long after occurrence of thefalling edge of the P_VGU signal.

The F_LATCH signal at an output node 223 of the compensation circuit 221is initially controlled by the COMP_(—)0 signal outputted by thecomparator 440. For each cycle of the SW voltage waveform beforeoccurrence of the falling edge of P_VGU, the COMP_(—)0 signal is highbecause the SW voltage is higher than V_(REF). A set state of the RSlatch 460 occurs when an A1 input of gate 464 goes low, which occurswhen P_VGU signal is high, such as before occurrence of the falling edgeof P_VGU. A high P_VGU signal is sufficient to define the set state ofthe RS latch 460 because, when either input of gate 464 goes low, theoutput of gate 464 goes high. Therefore, at the beginning of each cycle,the F_LATCH signal at the output node 223 of the compensation circuit221 is high.

Sometime after startup (during a first cycle), but while thecompensation circuit 221 is still in the initial state, a low COMP_(—)0signal occurs before a low COMP_AUX signal. For this reason, the stateof the F_LATCH signal at the output node 223 of the compensation circuit221 is initially controlled by a COMP_(—)0 signal of comparator 440 (seeCOMP_(—)0 to F_LATCH arrow in FIG. 5, Initial State). The low COMP_(—)0signal causes the output of gate 465 to go high. The high signaloutputted by gate 465 is fed into the A0 input of gate 464. A highP_VGU_B signal is fed into the A1 input of gate 464 (the P_VGU signal islow, and the inverter 450 causes the P_VGU_B signal to be high). As aresult, the output of gate 464 and the output of RS latch 460 go low.Therefore, the low COMP_(—)0 signal resets the RS latch 460, which meansthat the F_LATCH signal outputted by the compensation circuit 221 goeslow. Once the low COMP_(—)0 signal resets the RS latch 460, a highCOMP_(—)0 signal will not remove the RS latch from the reset statebecause the output of the RS latch is latched in the low state.

To remove the RS latch 460 from the reset state, i.e., to place the RSlatch in a set state, a low P_VGU_B signal is fed into the A1 input ofgate 464. This occurs when the P_VGU signal is high, and the inverter450 causes the P_VGU_B signal to be low. The set state of the RS latch460 results in the F_LATCH signal at the output node 223 of thecompensation circuit 221 being high. Therefore, a high P_VGU signalreleases the RS latch 460 so that it, and the compensation circuit 221,can accept another sample of the SW voltage.

Comparator 440 compares the SW voltage and the V_(REF) voltage.Comparator 440 sets a maximum time interval between occurrence of theSW=V_(REF) event (at such moment, upper power transistor 231 has alreadybeen turned off) and occurrence of the rising edge of the VGL signal(which causes lower power transistor 232 to turn on). Comparator 440ensures that the maximum time interval between the SW=V_(REF) event andoccurrence of the rising edge of the VGL signal is equal to a delaywithin comparator 440 plus a delay within the switcher logic 206 plus adelay within the SMPS 202. However, it is desirable that the timeinterval between the SW=V_(REF) event (when upper power transistor 231is already off) and the rising edge of the VGL signal (when lower powertransistor 232 turns on) be less than the delay within comparator 440plus the delay within the switcher logic 206 plus the delay within theSMPS 202. Therefore, other portions of the compensation circuit 221advantageously allow the time interval between the SW=V_(REF) event andthe rising edge of the VGL signal to be much shorter than the maximumdelay ensured by comparator 440. FIG. 5 illustrates that, in the initialstate, the time interval has a first duration 501.

The VGL signal is triggered by the F_LATCH signal. In one embodiment, arising edge of the VGL signal is triggered by a low F_LATCH signal aftera delay in the switcher logic 206. When the rising edge of the VGLsignal occurs (see F_LATCH to VGL arrow in FIG. 5, Initial State), thesampler circuit 401 samples the SW voltage. Because sampling occursafter the SW=V_(REF) event, when the compensation circuit 221 is in theinitial state, the sampled SW voltage at node 410 is below V_(REF). Thesampled SW voltage at node 410 pulls down current from inverted inputnode 409 of the amplifier 408 of the integrator 405. As a result, theVIN_AUX voltage starts to rise a little at each cycle.

Middle state. During a middle state, the falling edge of the P_VGUsignal starts the VC_AUX waveform, and the VIN_AUX voltage is now higherthan it was during the initial state. During the middle state, theVIN_AUX voltage is at such a voltage level that the VC_AUX=VIN_AUX event509 occurs before the SW=V_(REF) event. Therefore, the A1 input of gate465 goes low before the A0 input of the gate 465 goes low. Consequently,comparator 430 (and not comparator 440) controls the F_LATCH signal thatis outputted by the compensation circuit 221 (see VIN_AUX/VC_AUX toF_LATCH arrow in FIG. 5, Middle State).

During the middle state, the rising edge of the VGL signal causes thesampler circuit 401 to sample the SW voltage which, at such samplingmoment (see F_LATCH to VGL arrow in FIG. 5, Middle State), has a valuethat is lower than the V_(REF) voltage, and, as a result, the outputvoltage (VIN_AUX) of the integrator 405 continues to increase.

The time interval from the SW=V_(REF) event to the occurrence of therising edge of the VGL signal advantageously reduces (compared to theinitial state) as comparator 430 starts to control the F_LATCH signal atthe output node 223 of the compensation circuit 221. The time intervalfrom the SW=V_(REF) event to the occurrence of the rising edge of theVGL signal continues, advantageously, to reduce during successivesamples of the SW voltage while the compensation circuit 221 is in themiddle state. The switcher logic 206 generates a rising edge of the VGLsignal based upon the F_LATCH signal. For each cycle of the SW voltage,the moment that the rising edge of the VGL occurs approaches a littlecloser in time to the SW=V_(REF) event for the reason that the fallingedge of the F_LATCH signal continues to occur earlier relative to theSW=V_(REF) event. The falling edge of the F_LATCH signal continues toadvance in time for the reason that the VC_AUX=VIN_AUX event 509continues to advance with each cycle of the SW voltage. TheVC_AUX=VIN_AUX event 509 continues to advance in time for the reasonthat the output VIN_AUX of the integrator 405 continues to increase witheach cycle of the SW voltage. FIG. 5 illustrates that, in the middlestate, the time interval has a second duration 502 that is shorter thanthe first duration 501.

Synchronized state. After several cycles (in one embodiment, after 80cycles when a frequency of the SW voltage waveform is 8 MHz), thecompensation circuit 221 enters a synchronized state. The synchronizedstate is defined when a sufficient number of cycles have occurred suchthat the rising edge of the VGL signal occurs at the same moment thatthe SW voltage equals V_(REF). During the synchronized state, thefalling edge of the P_VGU signal starts the VC_AUX waveform. WhenVC_AUX=VIN_AUX, the comparator 430 is triggered, which generates veryshortly thereafter, a falling edge of the F_LATCH signal at the outputnode 223 of the compensation circuit 221 (see VIN_AUX/VC_AUX to F_LATCHarrow in FIG. 5, Synchronized State).

During the synchronized state, the rising edge of the VGL signal occursat about the same moment that the SW voltage equals V_(REF). FIG. 5illustrates that, in the synchronized state, the time interval has athird duration 503 that is advantageously very short and shorter thanthe second duration 502. The rising edge of the VGL signal is triggeredby the F_LATCH signal (see F_LATCH to VGL arrow in FIG. 5, SynchronizedState). The rising edge of the VGL signal causes the sampler circuit 401to sample the SW voltage. During the synchronized state, the SW voltageis falling through V_(REF). Therefore, during the synchronized state, avalue of the sampled SW voltage is equal to V_(REF). Consequently,during the synchronized state, the input voltage of the integrator 405becomes equal to V_(REF). Now, because both inputs of the integrator 405are equal to the V_(REF) voltage, the voltage (VIN_AUX) of the output ofthe integrator does not change, and a steady state is reached. Duringthe synchronized state, the F_LATCH signal outputted by the compensationcircuit 221 occurs, in one embodiment, 13 ns before occurrence of theSW=V_(REF) event. As a result, the driver stage 204 advantageouslyreduces the dead time, i.e., the interval of time when neither powertransistor 231 and 232 is on, to only about ten to a few hundredpicoseconds, in one embodiment. With the known driver stage 104, thedead time is disadvantageously much longer, e.g., about 10 ns to 40 ns.

The aforesaid explanation of the operation of the driver stage 204assumes that the delay from the P_VGL and P_VGU signals to the SWvoltage sweep, and the delay from the F_LATCH and R_LATCH signals to theVGL and VGU signals, respectively, do not change. In an actualimplementation of the device 200, any occasional changes in these delayscause the compensation circuit 221 to reach a new steady state tocompensate for the new values of the delays.

The driver stage 204 is designed such that the amount of time that theP_VGU signal occurs prior to occurrence of the SW=V_(REF) event is atleast 3 ns to 5 ns greater than the total delay for which thecompensation circuit 221 compensates. This is because the compensationcircuit 221 outputs the F_LATCH signal 3 ns to 5 ns after, and inresponse to, occurrence of the falling edge of the P_VGU signal.

The driver stage 204 advantageously can be used when the powertransistors 231 and 232 may be replaced by other power transistors thathave different characteristics, and/or when their delays are unknownand/or vary much, because any delay is compensated for by the TLL of thedriver stage 204. In this way, fast or slow power transistors 231 and232 can be used because, after a steady state is reached, all the delaysare compensated. A designer just need to know what should be a maximumexpected delay so that the P_VGU signal and the P_VGL signal can begenerated enough time in advance of the SW=V_(REF) event and theSW=V_(REFR) event, respectively, to compensate for a worst case delay.

FIG. 6 illustrates graphs 600 of results of operation of a simulation ofthe known device 100. The vertical axis of the top graph represents thevoltage SWA at node 133 of FIG. 1. The vertical axis of the bottom graphrepresents the voltage VGUA at the control electrode of upper powertransistor 131, and the voltage VGLA at the control electrode of lowerpower transistor 132 of FIG. 1. The horizontal axis represents a typical500 ns interval of time. FIG. 6 illustrates that a dead time 601 of along duration exists between the moment than the VGUA signal goes low(upper power transistor 131 turns off) and the moment that the VGLAsignal goes high (lower power transistor 132 turns on), and vice versa.FIG. 6 illustrates that the dead time 601 is approximately 0.02 μs=20ns. FIG. 6 also illustrates that the SWA voltage disadvantageously goesdown to −0.6V during each dead time. The SWA voltage going down to −0.6Vduring each dead time detrimentally effects efficiency of the known SMPS102. There are two factors have a detrimental effect on the efficiencyof the known SMPS 102: 1) the SWA voltage going down to −0.6V duringeach dead time, and 2) a long duration dead time of 20 ns. Both factors,when combined, generate an energy loss E_(LOSS)=0.6V×I_(R138)×20 ns,where I_(R138) is current through output resistor 138.

FIG. 7 illustrates graphs 700 of results of operation of a simulation ofone embodiment of the device 200 in accordance with the invention. Thevertical axis of the top graph represents the voltage SW at node 233 ofFIG. 2. The vertical axis of the bottom graph represents the voltage VGUat the control electrode of upper power transistor 231, and the voltageVGL at the control electrode of lower power transistor 232 of FIG. 2.The horizontal axis represents a typical 500 ns interval of time afterthe compensation circuit 221 has reached the synchronized state. FIG. 7illustrates that a dead time 701 of a very short duration exists betweeneach moment than the VGU signal goes low (upper power transistor 231turns off) and each moment that the VGL signal goes high (lower powertransistor 232 turns on), and vice versa. FIG. 7 illustrates that thedead time 701 is advantageously only approximately 0.1 ns=100 ρs. FIG. 7also illustrates that the SW voltage advantageously does not go down to−0.6V during each dead time. The SW voltage does not go down to −0.6Vduring each dead time because the VGL signal, which turns on the lowerpower transistor 132, occurs when the SW voltages reaches V_(REFR),which, in the simulation shown in FIG. 7, is −0.1V. As a result, thelower power transistor 132 turns on before the SW voltage has anopportunity to go down to −0.6V. The lower power transistor 132, whileit is on, prevents the SW voltage from going below −0.03V. In thesimulation results shown FIG. 7, the SW voltage advantageously does notgo below −0.2V.

FIG. 8 is a schematic diagram of one embodiment of circuit 402. Theembodiment of circuit 402 shown in FIG. 8 emulates the sampler circuit401 and the resistor 406 shown in FIG. 4. The embodiment of circuit 402shown in FIG. 8 comprises eight (8) switches S1 to S8, and capacitors801 and 802, which are configured as illustrated in FIG. 8. Capacitor802 is coupled between nodes 409 and 411. The embodiment of circuit 402shown in FIG. 8 also comprises a DC voltage source 803 that is coupledbetween node 411 and a ground terminal. In one embodiment, such DCvoltage is 1.0V. The embodiment of circuit 402 shown in FIG. 8 alsocomprises inverter 811 that receives the VGL signal and produces an OBsignal. Inverter 811 is coupled to a delay cell 812 that is coupled toan inverter 813 that produces an OD signal. Inverter 813 is coupled toan inverter 814 that produces an ODB signal. For each switch, S1 to S8,when the V>0, the switch is turned on, i.e., closed, and when V=0, theswitch is turned off, i.e., open, where V is a voltage controlling theswitch. The embodiment of circuit 402 shown in FIG. 8 continuallysequences through the following five passes. Each pass is defined by theVGL signal and the delay of delay cell 812. The VGL signal is controlledindirectly by the CLOCK signal.

Pass (1). When the VGL signal is low, the OD signal is low, and the OBand ODB signals are high. In this way, the switches S1, S5, S2 and S6are on, and the switches S3, S7, S4 and S8 are off. The capacitor 801 isconnected through switches between SW and V_(REF), and the voltage ofthis capacitor follows the waveform given by SW−V_(REF). This pass isdefined by a low state of the VGL signal.

Pass (2). At the rising edge of the VGL signal, the following switchingoccurs. VGL goes high and the OB signal goes low. This turns on switchesS3 and S7, and turns off switches S1 and S5. Now, capacitor 801 isfloating. The circuit of FIG. 8 stays in this pass while the delay cell812 maintains the low state.

Pass (3). After a delay of a few nanoseconds inserted by the delay cell812, the OD signal goes high and the ODB signal goes low. This turns onswitches S4 and S8, and turns off switches S2 and S6. Now, capacitor 801is in parallel with capacitor 802, and a charge of capacitor 801 istransferred to nodes 409 and 411. This pass is defined by a high stateof the VGL signal.

Pass (4). At the falling edge of the VGL signal, the following switchingoccurs. When the VGL signal goes low, the OB signal goes high. Thismeans that switches S3 and S4 are turned off, and switches S1 and S5 areturned on. Now, capacitor 801 is floating. The circuit of FIG. 8 staysin this pass while the delay cell 812 maintains the high state.

Pass (5). After a delay of a few nanoseconds inserted by the delay cell812, the OD signal goes low and the ODB signal goes high. This turns offswitches S4 and S8, and turns on switches S2 and S6. Now, capacitor 801is connected between the SW and V_(REF) nodes, and capacitor 801 followsthe waveform given by SW−V_(REF). This state is the same as in pass (1),and the sequence is complete. The sequence from pass (1) to pass (4) iscontrolled by the VGL signal.

For each pass, capacitor 801 follows the SW−V_(REF) voltage, and, at therising edge of the VGL signal, disconnects from the SW and V_(REF)nodes, and connects to nodes 409 and 411. When the rising edge of theVGL signal occurs, capacitor 801 is charged with a value of theSW−V_(REF) voltage that exists at the moment of the rising edge of theVGL signal.

Because capacitor 801 is switching with a frequency “f”, the charge ratethat this switching introduces into the inverted input node 409 ofamplifier 408 is equivalent to a current generated by resistor 406. Theresistance of resistor 406 is equal to (C₈₀₁ f)⁻¹, where C₈₀₁ iscapacitance of capacitor 801. The ideal sampler 401 is added because thesampled SW voltage is controlled by the VGL signal.

In FIG. 4, V_(REF) is shown connected directly to the input node 411 ofamplifier 408. On the other hand, in FIG. 8, V_(REF) is not connecteddirectly to the input node 411 of the amplifier 408. This is possiblebecause both nodes of capacitor 801 are switched, thereby preserving thecharge of capacitor 801, and the integrator 405 works as if V_(REF) wereconnected to the non-inverted node of the amplifier 408. In FIG. 8, theinput node 411 of amplifier 408 is coupled to the DC voltage source 803.The embodiment of circuit 402 illustrated in FIG. 8 is advantageousbecause the amplifier 408 can work with a more convenient bias voltagethan the V_(REF) voltage. The V_(REF) and V_(REFR) voltages (+0.1V and−0.1V, respectively) used in one embodiment of the driver stage 204 arevoltages that are difficult to use within an integrated circuit (becausethey are very close to 0V). If an integrated circuit is supplied by 3.3Vand 0V, such as in the embodiment of circuit 402 shown in FIG. 8, themore convenient bias voltage with which to work is in a range of 0.8V to2.5V (note that 2.5V=3.3V−0.8V). By using the embodiment of circuit 402shown in FIG. 8, the aforesaid difficulty with bias voltages can beavoided, and the compensation circuit 221 can consume less power andoccupy less area.

The driver stage 204 is not limited for use with a SMPS; the driverstage can be used with other circuits that have elements that areswitched on and off. For example, the driver stage 204 can be used withan H-bridge.

The device 200 is not limited to comprising the SMPS 202, and mayinstead comprise any circuit that includes two sub-circuits each ofwhich has a provision for being switched on and off. The upper powertransistor 231 is not limited to being a single transistor and mayinstead be a sub-circuit that has a provision for being switched on andoff. The lower power transistor 232 is not limited to being a singletransistor and may instead be another sub-circuit that has a provisionfor being switched on and off.

Although, in one embodiment, the switcher logic 206, the first delaycompensation circuit 221 and the second delay compensation circuit 222are disposed on an integrated circuit fabricated using CMOS technology,these circuits can also be disposed on an integrated circuit fabricatedusing other technologies. Although the invention has been described withrespect to specific conductivity types or polarity of potentials,skilled artisans appreciated that conductivity types and polarities ofpotentials may be reversed.

The specification and figures are to be regarded in an illustrativerather than a restrictive sense, and all such modifications are intendedto be included within the scope of the present invention. Any benefits,advantages or solutions to problems described herein with regard tospecific embodiments are not intended to be construed as a critical,required or essential feature or element of any or all the claims.Unless stated otherwise, terms such as “first” and “second” are used toarbitrarily distinguish between the elements such terms describe.

Thus, these terms are not necessarily intended to indicate temporal orother prioritization of such elements. Note that the term “couple” hasbeen used to denote that one or more additional elements may beinterposed between two elements that are coupled.

Although the invention is described herein with reference to specificembodiments, various modifications and changes can be made withoutdeparting from the scope of the present invention as set forth in theclaims below.

What is claimed is:
 1. A device, comprising: a circuit including atleast a first sub-circuit and a second sub-circuit, each of which havinga provision for being switched on and off, wherein the circuit producesa varying signal whose amplitude varies in response to the firstsub-circuit and the second sub-circuit being switched on and off; and adriver stage for controlling the circuit the driver stage including:switcher logic, coupled to the circuit, for producing a signal forcontrolling the first sub-circuit and for producing a signal forcontrolling the second sub-circuit, and a delay compensation circuitcoupled to the circuit and to the switcher logic, for generating atriggering signal for the switcher logic, wherein a changed state of thetriggering signal is aligned with a moment that an amplitude of thevarying signal equals a reference level, wherein the switcher logicproduces, in response to the changed state of the triggering signal, thesignal for controlling the first sub-circuit.
 2. The device of claim 1,wherein the driver stage includes: a second delay compensation circuitcoupled to the circuit and to the switcher logic for generating a secondtriggering signal for the switcher logic, wherein a changed state of thesecond triggering signal is aligned with a moment that an amplitude ofthe varying signal equals a second reference level, and wherein theswitcher logic produces, in response to the changed state of the secondtriggering signal, the signal for controlling the second sub-circuit. 3.The device of claim 2, wherein driver stage controls the circuit suchthat only one of the first sub-circuit and the second sub-circuit is onat any given instant of time.
 4. The device of claim 2, wherein thedriver stage is disposed on an integrated circuit.
 5. The device ofclaim 3, wherein the circuit is an output stage of a switched-mode powersupply, and wherein the first sub-circuit includes a first powertransistor and the second sub-circuit includes a second powertransistor.
 6. The device of claim 3, wherein the switcher logicproduces a changed state of a first prior signal before producing achanged state of the signal for controlling the first sub-circuit, andwherein the switcher logic produces a changed state of a second priorsignal before producing a changed state of the signal for controllingthe first sub-circuit.
 7. The device of claim 6, wherein the delaycompensation circuit generates the changed state of the triggeringsignal in response to occurrence of the changed state of the first priorsignal and in response to the amplitude of the varying signal.
 8. Thedevice of claim 6, wherein the second delay compensation circuitgenerates the changed state of the second triggering signal in responseto occurrence of the changed state of the second prior signal and inresponse to the amplitude of the varying signal.
 9. A switched-modepower supply (SMPS), comprising: a first power transistor and a secondpower transistor, wherein the SMPS produces an SW voltage whoseamplitude varies as a result of turning on and off of the first powertransistor and the second power transistor; and a driver stage, thedriver stage including: switcher logic, coupled a clock circuit, forproducing a first signal for timing turning on of the first powertransistor and a second signal for timing turning on of the second powertransistor during each cycle of a clock signal, wherein the switcherlogic produces a changed state of a third signal prior to producing achanged state of the first signal, and wherein the switcher logicproduces a changed state of a fourth signal prior to producing a changedstate of the second signal, a first delay compensation circuit, coupledto the switcher logic, for producing a first triggering signal inresponse to the changed state of the third signal and an amplitude ofthe SW voltage, wherein the first triggering signal causes the switcherlogic to produce the changed state of the fourth signal and the changedstate of the second signal, and a second delay compensation circuit,coupled to the switcher logic, for producing a second triggering signalin response to the changed state of the fourth signal and the amplitudeof the SW voltage, wherein the second triggering signal causes theswitcher logic to produce the changed state of the third signal and thechanged state of the first signal.
 10. The SMPS of claim 9, wherein thedriver stage is disposed on an integrated circuit.
 11. The SMPS of claim9, including: a first buffer, for producing a first turn-on signal fromthe first signal, wherein the first turn-on signal is coupled to acontrol electrode of the first power transistor, and a second buffer,for producing a second turn-on signal from the second signal, whereinthe second turn-on signal is coupled to a control electrode of thesecond power transistor.
 12. The SMPS of claim 11, wherein driver stagecontrols the SMPS such that only one of the first power transistor andthe second power transistor is on at any given instant of time.
 13. TheSMPS of claim 11, wherein the first delay compensation circuit producesa changed state of the first triggering signal in response to a momentthat a falling SW voltage passes through a first fixed DC referencevoltage.
 14. The SMPS of claim 11, wherein the second delay compensationcircuit produces a changed state of the second triggering signal inresponse to a moment that a falling SW voltage passes through a secondfixed DC reference voltage.
 15. The SMPS of claim 9, wherein the firstpower transistor includes a first MOSFET and the second power transistorincludes a second MOSFET, wherein a conducting electrode of the firstMOSFET and a conducting electrode of the second MOSFET are coupled to anode, and wherein the SMPS produces the SW voltage at the node.
 16. TheSMPS of claim 15, wherein the switcher logic aligns a moment of turningon of the first power transistor and a moment of turning on of thesecond power transistor with a moment that the SW voltage sweeps through0V.
 17. The SMPS of claim 13, in which the first delay compensationcircuit includes a sampler circuit that samples and holds a presentvalue of the SW voltage at each occurrence of a transition in the stateof the second turn-on signal.
 18. The SMPS of claim 17, in which thefirst delay compensation circuit includes an integrator, coupled to thesampler circuit, that integrates a voltage difference between a sampledSW voltage and the first fixed DC reference voltage, and outputs resultsof the integration, wherein a time constant of the integrator is greaterthan a period of the SW voltage.
 19. The SMPS of claim 18, in which thefirst delay compensation circuit includes: a waveform generator thatgenerates a decreasing voltage, and a comparator that has one inputterminal for receiving the decreasing voltage generated by the waveformgenerator, an inverted input terminal for receiving the results of theintegration of the integrator, and an output terminal for outputting asignal that generates the first triggering signal.
 20. An integratedcircuit, comprising: a delay compensation circuit, disposed on acircuit-supporting substrate, including: a terminal for receiving avarying signal, a sampler circuit that samples and holds a present valueof the varying signal at each occurrence of a transition in a digitalsignal, an integrator, coupled to the sampler circuit, that integrates avoltage difference between a sample of the varying signal and areference signal, and outputs results of the integration, wherein a timeconstant of the integrator is greater than a period of the varyingsignal, a waveform generator that generates a decreasing voltage inresponse to a transition in a second digital signal, and a comparatorthat has one input terminal for receiving the decreasing voltage, aninverted input terminal for receiving the results, and an outputterminal for outputting a signal that generates an output signal.